Fig. 1(a) and 1(b) show the planar and triangular array configurations of the Vivaldi antenna with slant polarization. For high-power jammer applications, the array antenna aperture needs to be as large as possible because the array requires high-gain properties. In addition, it is essential to determine the array distances (dx₁ and dy₁) and the total Vivaldi width (w) to avoid unwanted grating lobes for beam steering. In a uniform planar array, the array distances (dx′₁ = dy′₁ = 42 mm [0.84λ at 6 GHz]) along the x- and y-axes are equal. A grating lobe does not occur until the maximum steered beam at angles of Az = 12.2° and El = 11.9°. However, it did not achieve the required beam-steering angles (−45° ≤ Az ≤ 45° and −25° ≤ El ≤ 25°) in this research, as shown in Fig. 2(a). On the other hand, Fig. 2(b) presents the improved grating-lobe-free region of the triangular array, despite the array distance dx₁ being equal to that of the planar array. dx₂ (x-shifted array distance at an even-numbered row), and dy₁ are half of dx₁. These array distances result in an isosceles-right triangular array configuration that can increase the grating-lobe-free region up to the required beam steering region. Based on the results, the total Vivaldi width w can be determined as 29.7 mm, which is the square root of dx₁.

Fig. 3(a) and 3(b) illustrate the geometry of the proposed array element. The proposed Vivaldi element, based on Kindt’s model [9], is composed of radiating flares and a cavity part. The radiating flares have exponential curves using a function of f(z), as follows:

We determined the flare curves to achieve high directivity for high-power jammers. The length l₁ affects the number of resonances in the broadband frequency band. The cavity part is an important component of the proposed antenna geometry that can improve the broadband impedance matching characteristics. The total area of the metal component for the proposed element can be adjusted according to the width w₇ and length l₇ of the rectangular cavity. It can tune the antenna port reflections in the low-end frequency band because the electrical current path lengthens when the metal area increases. Then, the cavity and the radiating flares are connected through linear transmission lines, which enable fine-tuning of the broadband impedance matching characteristics. Fig. 3(c) presents the extension of the periodic structure for observing the AVSWR in consideration of dual-slant polarization. Vivaldi array elements are rotated by ±45° and integrated so that their edge sides are shared. Herein, the width w₂ is half the Vivaldi thickness t. We then optimize the proposed array element parameters based on the infinite periodic array to achieve broad frequency band operation, and the detailed parameters are listed in Table 1. Fig. 4 shows the simulated AVSWRs of the periodic structure from 2–6 GHz at steering angles (Az, El) of (0°, 0°), (45°, 0°), and (0°, 25°). There is an abnormal peak near the central frequency band that can be adjusted by the parameters t and l₁. In our research, the thickness should be maximized to enhance the array gain in the low-end frequency band, so that t and l₁ are optimized by 10 mm and 56 mm, respectively. to achieve the maximum AVSWRs at steering angles (Az, El) of (0°, 0°) and (45°, 0°) below 3:1 over the operating frequency band.

Fig. 5 represents the proposed array antenna having dual-slant polarization with 32 elements (a 4 × 4 triangular array configuration) for high-power jamming applications. To observe the array performance, the +45° slant-polarized elements are excited to obtain the S-parameter matrix and active element patterns, while the other polarized elements are terminated with 50-Ω loads. Fig. 6(a) shows each component of the proposed array antenna: radiators, a cavity part, a ground plate, and SMA feeders. In particular, the modularized radiating components can easily be replaced in case of breakdown or malfunction. Fig. 6(b) illustrates the assembly process employing bolts and nuts for high durability and easy fabrication. Fig. 6(c) and 6(d) show photographs of the fabricated array antenna.

To observe the AVSWRs of the proposed array, the +45° slant-polarized elements are measured using a two-port vector network analyzer. All ports are measured to obtain a 16 × 16 S-matrix for the AVSWR calculation according to the Az and El directions as follows [10–12]:

Γ_i(Az,El) is the active reflection coefficient of the i-th port according to the steering angle in the azimuth and elevation directions, and i is the port index number (i =1, 2, …, I). m and n are the index numbers of column and row elements (M = 1, 2, 3…, and N = 2, 4, 6 …). Fig. 7 shows the AVSWR results at steering angles (Az, El) of (0°, 0°), (45°, 0°), and (0°, 25°) for Port 1 (the edge element) and Port 6 (the center element). The measured and simulated AVSWRs for Port 1 are below 2.74:1 and 2.99:1 from 2.5 GHz to 5.5 GHz. The peak measured and simulated AVSWRs for Port 6 are 2.85 and 2.88, respectively, and these results agree well with each other. In addition, we obtain the array mismatch efficiency of the proposed array antenna as a figure of merit to evaluate the array performance for the high-power jammers using Eq. (6) [13].

Fig. 8 illustrates the measured and simulated array mismatch efficiency of the proposed array antenna at steering angles according to the frequency. The measured averaged array mismatch efficiencies over the entire frequency range are 91.3% (Az = 0°, El = 0°), 87.2% (Az = 45°, El = 0°), and 90.9% (Az = 0°, El = 25°), and those of the simulations are 93.6%, 90.4%, and 93.4%, respectively.

The AEPs for all ports of the proposed array are measured in an anechoic chamber to obtain the array radiation performance. The array gain is calculated using the AEPs of all ports based on Eq. (7) [14, 15]:

where g_i is a complex AEP vector of the i-th port, and w_i is a complex weighting vector of the i-th port. Fig. 9 shows comparisons of the measured, simulated, and theoretical array gains in the bore-sight direction. The theoretical result is calculated as 4πA/λ², where A is the aperture area of the proposed array (in this case, 179 cm²). All the results agree well with each other, and the measured array gain is above 9.5 dBi from 2 GHz to 6 GHz. To examine the beam steering performance, the complex weighting vector w_i for each port is obtained at a high-end frequency of 6 GHz in accordance with the steering angle in the azimuth and elevation directions. Fig. 10 shows the beam steering performances along the E- and H-planes according to the steering azimuthal (Az = 0°, 15°, 30°, 45°) and elevational (El = 0°, 10°, 20°, 30°) angles at 6 GHz. The measured peak side lobe levels along the azimuth and elevation directions are 10.2 and 10.7 dB, respectively. The steered beam patterns can maintain high gain properties without the existence of grating lobes. The proposed array antenna performances are compared with previous works, as shown in Table 2 [6, 9, 16–19], for further explanation.