As wireless mobile communication develops rapidly and continuously, the radio spectrum is being reduced, and frequency is becoming a precious resource [1]. Multiple-in multiple-out (MIMO) wireless transmission technology, which applies multiple antennas, can increase the wireless transmission capacity without requiring extra spectrum resources [2, 3]. Additionally, owing to the diversity gain by MIMO, a relatively small transmitter power is required for each transmission antenna. Therefore, an expensive amplifier is not required.

As the number of indoor and urban wireless access points increases, the signal-to-noise ratio decreases, resulting in a multipath environment in which signals become distorted. Destructive interference from multipath radio waves increases the vulnerability of wireless access points. Diversity technology is one way of solving this problem [4, 5].

A pattern-diversity antenna can reduce blind spots by implementing radiation patterns with different null positions simultaneously [6]. Therefore, noisy environments can be avoided, and environmental adaptability can be increased. Additionally, the system energy can be reduced by appropriately adjusting the direction of the signal to the target, and the coverage can be increased by redirecting the direction of the main beam.

A common method of implementing a dual-feed pattern diversity antenna is to place two identical antennas at a certain distance such that radiation occurs in different directions [7–10]. Many decoupling structures have been proposed to obtain good isolation characteristics between two antennas [7, 8, 11]. Because this structure uses the same antenna, it exhibits the same broadside radiation pattern. Additionally, it is easy to increase the number of antennas for MIMO applications. Consequently, the channel capacity increases. In [12] and [13], two antennas of different shapes were used to obtain broadside and conical radiation patterns. In this case, it is important to obtain good isolation characteristics even while designing an extremely short separation distance between the two antennas to prevent an increase in the size of the entire antenna system.

In contrast, a pattern diversity antenna with broadside and conical radiation patterns based on a common radiator has also been proposed. In this radiator, the mode generated in the resonator varies according to the feed position; thus, different radiation patterns can be obtained. The first type is an antenna with two input ports. The two feed points were designed differently to have broadside and conical radiation patterns [14, 15]. In the second type, a hybrid coupler is used as the feed structure. Hybrid couplers exhibit good reflection coefficients and isolation characteristics over a wide bandwidth range. Therefore, an antenna with a hybrid coupler has a wide bandwidth [16]. In [17], broadside and conical radiation patterns were implemented by adjusting the phase difference applied to the antenna using a power divider combined with a radio frequency (RF) switch. This structure does not consider isolation because it has only one input port; however, there are disadvantages, such as distortion of the radiation pattern owing to the bias line, performance degradation, and DC power consumption. In Table 1, the performances of previous polarization diversity antennas and the proposed antenna are compared.

The proposed antenna consists of two semi-circular slots and a modified 90° hybrid coupler. The designed antenna uses a substrate (TLY-5) with a permittivity of 2.2 and a thickness of 31 mil. Two semi-circular slot antennas were implemented by placing a stripline of width W at the center of a circular slot of radius R. The slot antenna and ground plane were implemented on the backside of the substrate and correspond to the gray area in Fig. 1. A 90° hybrid coupler was implemented on the top of the substrate, which corresponds to the orange area. It is well known that signals with the same magnitude and a phase difference of 90° arrive at the two output ports of a conventional hybrid coupler.

In other words, when the RF power was incident on Port 1, the phase of the signal at point B was 90° greater than that at point A. As the 90° delay line in the shaded area was added, the phase of the BB′ line became 90° longer than the phase of the AA′ line; therefore, the phase of the signal at point B′ was 180° greater than that at point A′. At points A′ and B′, the RF power was coupled to a slot of width w in the hybrid coupler. The phase difference applied to the lower and upper semi-circular slots was 180°. Accordingly, a broadside radiation pattern was formed by synthesizing the power radiated by the two semicircular slot antennas.

When the RF power is incident on Port 2, the phase of the signal reaching point A on the lower line is 90° greater than that of the signal reaching point B on the upper line. Owing to the 90° delay line on the upper line, the phases of the signals arriving at points A′ and B′ are the same. Accordingly, signals with the same phase were applied to the two semi-circular slots to form a conical radiation pattern. The coupler used is composed of 50 Ω lines and 35.35 Ω lines in the same manner as the conventional 90° hybrid coupler. The input and output ports are set to 2.4 mm, corresponding to 50 Ω of the substrate used. The simulation was performed using Ansys High-Frequency Structure Simulator (HFSS). Fig. 1(b) shows a photograph of the fabricated antenna.

As mentioned above, the proposed structure consists of a slot resonator that acts as a radiator and a deformed coupler that serves as a feeding structure. The frequency responses of the radiator and feed structures are shown in Figs. 2 and 3. Fig. 2(a) shows the single-feed antenna while the shape of the radiator is maintained, and Fig. 2(b) shows the dual-feed antenna without the use of a hybrid coupler. The resonant frequencies of the single- and dual-feed antennas were identical. The dual-feed antenna without a hybrid coupler exhibited poor isolation characteristics. This is because the two semi-circular slots are located close to each other.

Fig. 3 shows the coupler used in this study. The coupler in Fig. 3(a) was implemented such that the phase difference at the output stage was 180° by adding a 90° delay line to the conventional 90° hybrid coupler. As shown in the simulation results, the operating frequency of the coupler was designed to be the same as the resonant frequency of the radiator, and the phase difference between Ports 3 and 4 (output stage) was 180° at the center frequency. In terms of |S₁₁| < −15 dB, at the lower edge, S₂₁ = −2.73 dB, S₃₁ = −4.05 dB, and the phase difference at this time is 165°. Meanwhile, at the upper edge, S₂₁ = −3.38 dB, S₃₁ = −3.15 dB, and the phase difference is 184°.

Fig. 4 shows the current distribution in the antenna according to the input port. The simulation was performed at 3.5 GHz, and the parameters of the simulated antenna were R = 23 mm, W = 4.3 mm, and w_s = 0.7 mm. The red and blue areas indicate strong and weak currents, respectively. In Fig. 4(a), when power is applied to Port 1, the phase difference between the two output ports of the conventional hybrid coupler is 90°. However, when a 90° delay line is added to the right output port, the phase difference between the left and right output ports is 180°.

The current coupled to the narrow slit flows equally to the left on both sides. Consequently, a current distribution is generated such that the left and right semi-circular slots have a phase difference of 180°. Because the half-circular slot antennas are in opposite directions, the current distribution forms a broad-side radiation pattern. From the perspective of the current distribution formed on the stripline located at the center of the circular slot, the currents forming from top to bottom and bottom to top coexist, resulting in current cancellation. This explains the good isolation characteristics between the left- and right-slot antennas.

Fig. 4(b) shows the current distribution when power is applied to Port 2. The two output ports of the conventional hybrid coupler have a 90° phase difference; however, when a 90° delay line is added, both output ports have the same phase. The direction of the current coupled to the narrow slit was reversed. Consequently, a current distribution is formed such that the left and right semicircular slot antennas are in the same phase. Because the semicircular slot antennas are in opposite directions, the current distribution forms a conical radiation pattern. From the perspective of the current distribution formed in the middle stripline, the currents flowing from top to bottom and bottom to top were formed simultaneously, resulting in an offset effect. This explains the good isolation characteristics between the left- and right-slot antennas.

Fig. 5 shows the equivalent circuit model of two half-circular slot antenna pairs. In the radiating element circuit model, C_a, L_a, and R_a represent the capacitance, inductance, and radiation resistance of the half-circular slot antenna, respectively. In the coupling circuit model, C_c represents the coupling effect between the microstrip and the slot, and C_s, L_s, and R_s represent the capacitive effect, inductive effect, and loss in the substrate, respectively. When the circuit is excited by CM or DM signals, the center symmetry plane can be equivalent to perfect magnetic conductor (PMC) or perfect electric conductor (PEC), respectively. Therefore, in the equivalent circuit model, the resonant frequencies of CM and DM are as follows:

Fig. 6(a) shows the simulation results of the reflection coefficients of the two ports and isolation between the two antennas according to the change in the radius R of the circle. All the antenna design parameters were kept the same except R, which was the same as mentioned above. As R increased, the frequency response of Port 1 decreased slightly; however, the change was not significant. However, as R increased, the frequency response of S₂₂ decreased relatively rapidly.

The frequency range in which S₁₁ is less than −10 dB has broadband characteristics, whereas the frequency range in which S₂₂ is less than −10 dB is very narrow. Therefore, the frequency band in which the polarization diversity antenna operates depends on S₂₂. R = 23 mm was selected due to the matching characteristics of S₂₂, the correspondence between the resonant frequency of Port 2, and the null frequency of S₂₁. At this time, S₂₁ in the operating band is lower than −20 dB.

Fig. 6(b) shows the simulation results according to the change in width W of the stripline dividing the circular slot by half. The antenna design parameters are identical to those described above. W is a parameter that affects the matching characteristics of S₁₁. At this time, the isolation hardly changed. Considering the reflection coefficient characteristics in Port 1, W was selected to be 4.3 mm.

Fig. 7 shows the simulation results according to the change in l_s. The antenna design parameters are identical to those described above. As l_s increased, the resonant frequencies of Ports 1 and 2 tended to decrease. At this time, S₂₁ hardly changed. As can be seen from the simulation results, the frequency response of S₁₁ shows a wideband characteristic, while the frequency response of S₂₂ shows a narrowband characteristic, and as l_s changes, the overlapping bandwidth barely changes at about 8%.

To match the characteristics of S₁₁, l_s should be 16.2 mm or 17.2 mm. However, considering that the characteristics of S₂₁ are best around 3.5 GHz and that the resonant frequencies of S₁₁ and S₂₂ are near 3.5 GHz, l_s was selected at 17.2 mm.

Fig. 8 shows the simulation results according to the change in l_stub. The antenna design parameters are identical to those described above. As l_stub changes, the change in the frequency response of S₁₁ is relatively larger than that of S₂₂. Moreover, as l_stub increases, the overlapping bandwidth remains almost unchanged. Considering the matching characteristics of S₁₁ and S₂₂, it was best when l_stub was 13 mm. Even though the change in l_stub was set to 4 mm, the change in isolation characteristics was very small.

A prototype of the proposed two-port antenna was fabricated to validate the simulation results. Fig. 9(a) shows the simulated and measured frequency responses (S₁₁, S₂₁, and S₂₂) of the proposed antenna. The measured results agreed well with the simulation results. The discrepancy observed between the simulated and measured S-parameters can be attributed to fabrication tolerances. The measured results show impedance bandwidths of 1,245 MHz (2.64–3.885 GHz) and 290 MHz (3.315–3.605 GHz) for Ports 1 and 2, respectively. Thus, an overlapping impedance bandwidth of 290 MHz (3.315–3.605 GHz) was obtained for the two-port antenna system. As shown in Fig. 9(a), the isolation between the two antennas in the entire band is better than 30 dB, indicating that the two antennas are integrated with little mutual disruption.

The normalized simulated and measured far-field radiation patterns corresponding to the excitation of each port in the xz-and yz-planes are shown in Fig. 9(b) and 9(c), respectively. Only one port was measured at a time, and the other port was terminated by a 50 Ω load. The radiation patterns of the two beams are complementary. The discrepancies observed between the simulated and measured radiation patterns can be attributed to fabrication tolerances. Fig. 9(b) shows the E- and H-plane radiation patterns of Port 1. As shown in Fig. 9(b), a unidirectional radiation mode was observed as expected. The co-polar fields were found to be much stronger than the cross-polar fields, which was desirable.

Fig. 9(c) shows the E- and H-plane radiation patterns of Port 2. The radiation patterns in the xz- and yz-planes are conical, similar to those of a monopole antenna. The measured gain at Port 1 is 5.47 dBi. For Port 2, the measured gain was 3.64 dBi. The cross-polarization level is −14 dB when Port 1 is applied and −12 dB when Port 1 is applied. Fig. 10 shows the simulated and measured gain.

In this study, a two-feed pattern diversity antenna was designed, fabricated, and tested. The proposed structure consists of a circular slot resonator and a modified hybrid coupler with a 90° delay line. The circular slot was divided into two semicircular slots using a protruding stub that acted as a separate antenna. Based on the analysis of the current and electric characteristics, the odd and even modes of the circular slot were separately excited for the two radiating mode types. Consequently, a broadside or conical radiation pattern was obtained. The isolation between the two antennas exceeded 30 dB. The results indicated that the proposed design can provide MIMO with diverse characteristics.