Wideband Transmitarray Antenna Based on Two-Layer Aperture-Coupled Metasurface Unit Cell
Article information
Abstract
In this paper, a transmitarray antenna is designed and fabricated based on a two-layer aperture-coupled patch unit cell. The unit cell is composed of two layers of dielectric substrate with a simple narrow rectangular aperture between the layers. Through characteristic mode analysis, it is shown that while a single patch on each layer does not provide the 360° phase range required in the transmitarray antenna design, if the same patch is divided into several sub-patches, named metasurface, the 360° phase range is achieved. It is also shown that metasurface sub-patches provide a wide gain bandwidth and high aperture efficiency for transmitarray antennas. In addition, a 13 × 13 metasurface-based transmitarray antenna is designed and fabricated. When using a 3 × 3 metasurface unit cell, the measured −1 dB gain bandwidth is 19% and its measured gain is 25.7 dB at the center frequency of 10.5 GHz, exhibiting 50% overall aperture efficiency.
I. Introduction
High-gain antennas are a crucial requirement in various applications, including microwave, satellite communication, wireless system, imaging, and radar applications. The preferred antenna for meeting these diverse demands is the parabolic reflector antenna. To eliminate the complexity involved in manufacturing such antennas, planar reflectors, also known as reflectarrays, fabricated using printed circuit technology were introduced. Furthermore, to address feed blockage issues, planar transmitarray antennas with similar functions were introduced in the transmission mode. These antennas have attracted significant interest in recent years. Notably, a planar transmitarray antenna comprises multiple unit cells, each contributing a phase shift to the wave front, thereby compensating for the spatial phase delay from the feed to each element. In this context, it is integral to use unit cells capable of delivering around 360° of phase shift. Different types of unit cells have been proposed in the literature to enable proper phase shifts in transmitarray antennas while also maintaining a high transmittance value. The most widely employed unit cell for this purpose is the multilayer frequency-selective surface, which is employed to design transmitarrays with a 360° phase range.
In [1], a three-layer unit cell comprising a cross dipole, square loop, and cross slot is discussed. The researchers determined that the maximum achievable transmission phase range for −1 dB transmission depends on a number of factors, including the number of layers, substrate material, and layer separations. Interestingly, it was reported that the use of four layers can achieve a complete phase range of 360° for a −1 dB transmission coefficient, regardless of the shape of the elements.
Another type of unit cell that does not employ dielectric substrates, instead using only metallic layers and air to create a 360° phase shift is proposed in [2–4]. In [2], the bandwidth corresponding to a 3 dB gain reduction was 12%, while the aperture efficiency of the array was 24%. In [3], an aperture efficiency of 55% was obtained owing to the use of a suitable feed horn, along with the mitigation of dielectric loss and a −1 dB gain bandwidth of 15.5%. In [4], the proposed unit cell was characterized by a four-layer double split-ring slot structure in the Ku-band. The aperture efficiency and −1 dB gain bandwidth obtained were 55% and 7.4%, respectively. However, these unit cell structures did not exhibit sufficient endurance—they were easily deformed by the smallest pressure—which made the results of the array antenna susceptible to change.
Furthermore, researchers have recommended a type of transmitarray based on multilayer unit cells, where the layers are different and independent in form. For instance, in [5], a 360° phase shift range was achieved using a unit cell with three conducting layers, indicating that even in the case of a unit cell with a total electrical thickness of λ0/4, maximum amplitude of transmission coefficient can be achieved. In [6], the unit cell comprised two dielectric layers, with circular ring slots on their outer surfaces and an inner layer featuring a uniplanar compact photonic bandgap (UC-PBG) element. The inner layer served as both an extra resonator and a coupler. The overall thickness of this unit cell was 0.033λ0. At a frequency of 9.7 GHz, −1 dB gain bandwidth and aperture efficiency of 5.7% and 38% were reported, respectively. Meanwhile, authors of [7] presented a hybrid design concept featuring frequency selective surface and patch antenna elements in the same aperture. At the central frequency of 13.3 GHz, with a total thickness of 0.07λ0, −1 dB gain bandwidth of 6.7% and an aperture efficiency of 30% were attained.
Since its introduction in 1985 [8], aperture coupled microstrip antennas have proven to be valuable in various applications. The versatility and flexibility of its fundamental design have prompted extensive development and design variations worldwide [9]. Recent advancements and beneficial features of these antennas include impressive impedance bandwidths ranging from 5% to 50%, the ability to independently select antenna and feed substrate materials, the potential for theoretically zero cross-polarization in principle planes, and numerous options for patch shape, aperture shape, feed line type, radomes, and more [9].
Aperture coupled structures have been used to design reflectarray unit cells [10–12] and, recently, transmitarray antennas [13, 14]. For instance, a reflectarray based on a planar aperture coupled antenna configuration is presented in [10]. The required phase shift at each position of the reflectarray surface was obtained through changing in the slot length on the ground plane. In [12], the bandwidth characteristics of aperture coupled reflectarrays were examined by considering phase curves as a function of the operating frequency for various lengths of a tuning stub. However, these unit cells presented the reflectarrays with a limited operating bandwidth. In [13], a transmitarray antenna characterized by 6 substrates and aperture coupling between the layers employed time delay lines to provide the phase shift, as opposed to changing element dimensions, to achieve phase compensation. The antenna array achieved an efficiency of 50% and a −1 dB gain bandwidth of 17.6%. In [14], a 140-GHz low-temperature co-fired ceramic-integrated transmitarray antenna that employs wideband substrate-integrated waveguide aperture-coupling phase delay is proposed. This structure employed three substrate layers to provide 6.9% −1 dB gain bandwidth with 44% efficiency.
There has been growing interest in metasurfaces (MS) due to their flexibility and ease of manufacturing, which has led to their application in various fields. In particular, two-dimensional MS composed of small, electric surface scatters has gained immense popularity in antenna design, since it allows for improved performance and reduced antenna size. In [15], it was observed that an MS positioned directly above or below a feeding aperture or microstrip patch can greatly enhance impedance bandwidth and antenna gain.
However, the process of designing antennas based on the scattering parameters of incoming plane waves is often intricate. Therefore, our objective is to discover a more straightforward approach to designing MS structures. In this context, by employing the characteristic mode analysis (CMA) technique to examine various conductors, the surface of MS structure can be broken down into a sequence of mode currents that are mutually independent. These currents are unaffected by other factors, such as the position of the feed, and depend solely on the properties of the conductor itself, such as its size and shape. As a result, this analysis method not only ensures precise evaluations but also facilitates visualization of the antenna’s resonant characteristics [16].
Unlike the complicated designs proposed in the literature [13, 14], this paper presents the design of a simple two-layer aperture coupled patch as a unit cell for transmitarray antennas. Using a square patch as a unit cell does not provide the required phase shift range for transmitarray elements. To address this, through the use of CMA, the simple patch is divided into smaller patches, referred to as metasurface sub-patches, and used as the new unit cell configuration for the array. By adjusting the dimension of the MS sub-patches along with the size of the aperture coupling, the proposed unit cell is able to achieve the required 360° phase shift. Full-wave simulations of the unit cell and the transmitarray antenna are conducted, showing a −1 dB gain bandwidth of 19% and peak efficiency of 50% (whereas the single patch unit cell delivered only 7% gain bandwidth). The antenna is also fabricated and tested, and the results are presented.
II. Unit Cell Design
It is well known that a simple microstrip patch antenna, if divided into several smaller patches, can provide a wider impedance bandwidth [17]. This is because when a patch is subdivided into many smaller patches, many closely spaced resonances take place, which collectively appear as larger bandwidth resonance [18].
In this section, the above idea is relied on to design the proposed unit cell for the transmitarray antenna. Through CMA, a single patch is altered into MS sub-patches capable of achieving the required 360° phase range and a higher −1 dB gain bandwidth.
1. Slot Coupled Patch
In this study, an aperture coupled unit cell for a transmitarray antenna is presented. The geometry of the unit cell is illustrated in Fig. 1. It comprises transmitting and receiving MSs—elements that are placed on the top and bottom layers respectively—and the aperture coupling between these layers is a simple rectangular slot. The dielectric substrate is Rogers RO4003C, with a dielectric constant 3.55, loss tangent 0.0027, and thickness h = 0.813 mm. The MS design process is depicted in Fig. 2. The simplest element that can be used for transmitting and receiving elements is a single patch. This configuration is presented in Fig. 2(a) as Configuration A.
Proposed aperture coupled MS unit cell.
A unit cell with a fixed size of Wg × Wg depicting the evolution of MS sub-patches with different Ns: (a) conventional single patch (N = 1), (b) MS patch (N = 2), (c) MS patch (N = 3), and (d) MS patch (N = 4).
2. Design of the MS
To increase the phase range of the proposed unit cell, the single patch was replaced with MS sub-patches to create additional modes for the resonances of the unit cell and improve the S21 bandwidth, as will be explained in the next section.
The evolution of the MS is outlined in Fig. 2. The square patch was divided into multiple sub-patches with different Ns (=1, 2, 3, 4). The width of the unit cell for all Ns is considered Wg. The spacing between adjacent patches is referred to as Wsp, and the width of the sub-patches is given by Wp = [W − (N − 1) Wsp]/N. The width and length of the slot on the ground plane are expressed as Ls and Ws, respectively. The design parameters of the unit cell for the proposed transmitarray antenna are provided in Table 1.
Unit cell dimensions
Table 2 shows the relationship between MS size and resonant frequency. For N = 1, a single patch with a dimension of 0.44λ0 provides a resonant frequency of 0.032λ0, while for N = 2, featuring two smaller patches, the MS with a dimension of 0.76λ0 provides a resonant frequency of 0.032λ0. Therefore, to increase N so that it has almost the same resonant frequency as the other Ns, the overall size of the MS needs to be increased.
Summary of resonant frequency and overall MS size
Notably, in a later section, the size of the MS will need to be changed to attain the required phase shift (360°) for the array, since configuration N ≥ 4 will become too large to be held within the fixed width of the unit cell Wg.
3. Characteristic Mode Analysis of a Single Patch and MS Sub-patches
Recently, characteristic mode (CM) theory was employed to forecast MS characteristics, with results showing its significant potential for comprehending and analyzing the physical properties of antennas based on MSs [19].
To examine the functioning of the proposed MS, its modal characteristics were first examined in the CST Studio software package. Details regarding the evolution of the MS and the corresponding resonant frequency of the fundamental broadside mode are shown in Fig. 2.
Due to CST-related constraints, an ideal lossless substrate was utilized to simulate the CM, with the investigation focusing on a single patch and MS patches without incorporating a feeding structure. The desired mode resonances were achieved by modifying the dimensions and placement of the sub-patches within the MS configuration. The modal significance of the single patch and MS sub-patches are shown in Fig. 3. Four notable modes were detected within the frequency range, with Configuration A exhibiting a narrow band and Configurations B and C displaying wideband characteristics.
Modal significance of a single patch in Configuration A (a), and MS sub-patches in Configuration B (b) and Configuration C (c).
To further explain the radiating modes, Fig. 4 illustrates the respective radiation patterns associated with each mode. Modes 1 and 2 exhibit a broadside radiation pattern, while Modes 3 and 4 demonstrate a null in the broadside direction. Based on the results in Figs. 3 and 4, one can easily conclude that Modes 1 and 2 are preferable in Configurations B and C.
Modal patterns of the single patch in Configuration A (a), MS Configuration B (b), and MS Configuration C (c).
Therefore, if an x-plane wave is incident on any of the configurations, Mode 1 will be excited due to narrow slot placed along the x-axis. Furthermore, if a y-directed plane wave is incident on the narrow slot placed along the y-axis, Mode 2 will be excited. Since the slot is placed along the x-axis in this study, only Mode 1 will be effective. In addition, cross-polarization would be reduced.
4. Floquet Port Analysis of Unit Cells
Using the modal analysis discussed in the preceding section as the basis, a multiresonance unit cell with a wide frequency range is designed by employing MS patches. The structure of the unit cell is shown in Fig. 1.
A transmitarray antenna is usually made up of units arranged in a periodic manner. Full wave simulation software (High-Frequency Structure Simulator [HFSS]) utilizes periodic boundary conditions to account for the interaction between these units. To simulate the behavior of the unit cell with x-polarization (according to slot polarization on the ground plane), the TM mode was applied to the port.
To understand the behavior of the proposed structure, a simple aperture coupled patch structure as transmitarray unit cell was analyzed (Configuration A) and then compared to an aperture coupled MS sub-patch structure (Configurations B and C) as transmitarray unit cell. Furthermore, to investigate the resonant characteristics of the proposed MS unit cell, its scattered parameters were analyzed using the Floquet-port HFSS model.
The phase and amplitude of the transmission S21 for the single patch unit cell—Configuration A—was simulated, the results of which are shown in Fig. 5. It is evident that the single patch has a limited phase range of about 240°. As such, some elements will inevitably have unattainable phase shifts, owing to which the radiation patterns of these elements show increased sidelobe levels (SLLs), accompanied by a reduction in antenna gain [20].
Fig. 5 also presents the simulated phase and amplitude of S21 for MS unit cell Configurations B and C, based on the parameter values in Table 1. It is observed that the MS unit cells achieved better phase slopes than Configuration A.
The phase ranges of MS unit cell Configurations B and C were about 360°, which is higher than the 240° of the single patch unit cell in Configuration A. Notably, due to the slope of phase variation in Configuration C, it was chosen as the preferred unit cell for designing the transmitarray antenna.
To create Configuration C from Configuration A, apart from subdividing the patch into several smaller patches and adjusting the size Wp of the sub-patches, the length of the rectangular aperture also had to be adjusted according to Ls = Wp + 0.25 mm to achieve the full 360° transmission phase range using one unit cell. Fig. 6 illustrates the transmission magnitude and phase of the proposed MS unit cell for different sub-patch dimensions Wp. It is observed that a 360° phase shift is obtained when Wp changes from 2.5 to 5 mm.
Simulated transmission coefficients of Configuration C MS aperture coupled unit cell for various Wp: (a) magnitude and (b) phase.
Fig. 7 shows the transmission magnitude and phase of the proposed MS unit cell for different sub-patch dimensions, Wp, at various frequencies. The phase responses at the frequencies exhibit nearly parallel behavior, highlighting the potential for achieving a transmitarray antenna with wideband capabilities using the proposed unit cells
Simulated transmission coefficients of the Configuration C MS aperture coupled unit cell for various Wps across various frequencies: (a) magnitude and (b) phase.
Fig. 8 traces the variation in both magnitude and phase of transmission for the Configuration C unit cell when subjected to different incident angles. It is observed that, up to a 30° incident angle, the measured insertion loss is less than −1.8 dB. Furthermore, the phase curves of the MS unit cell remain parallel across various incident angles ranging from the center-placed patch (θ = 0°) to the corner-placed patch (θ = 30°) for F/D = 0.8, which was therefore used in the transmitarray design.
Simulated transmission coefficients of the MS unit cell (Configuration C) for various incident angles θ0: (a) magnitude and (b) phase.
III. Transmitarray Antenna Design
In this section, MS unit cell Configuration C is utilized as the fundamental component for constructing a wideband transmitarray antenna. A 13 × 13 transmitarray antenna is considered for this purpose. Furthermore, the feed antenna employed is an X-band corrugated conical horn antenna, as outlined in [21]. The horn antenna exhibited a stable gain over the entire bandwidth of 2.2 GHz. It was positioned in alignment with the transmitarray’s central axis at a distance of 153.6 mm or 5.12λ0 (F/D = 0.8) from the array. As mentioned before, while a simple conventional patch has a resonant frequency mode, MSs exhibit wider mode behavior, and the slot itself resonates as well. As a result, an MS patch unit cell can be expected to display a wider frequency response behavior and to achieve an extended phase range compared to a simple aperture coupled patch unit cell.
In this context, to ensure the effectiveness of a specific mode (Mode 1 or Mode 2, as mentioned earlier), it is crucial to appropriately design the transmitarray antenna.
This involves careful selection and excitation of the desired mode while preventing the activation of undesired modes. In other words, to achieve a high gain in the desired direction, it is important to establish well-matched impedance through the feeding mechanism, specifically targeting the symmetric mode while avoiding any excitation of the antisymmetric mode [22]. Therefore, a rectangular aperture is used as a coupling mechanism between the receive and transmit patches to maximize transmission efficiency and excite the fundamental mode at broadside. Finally, by properly adjusting the width of the MS sub-patches, Wp, and the length of the coupled slot, LS, a proper phase shift can be achieved.
1. Effect of Limited Phase Range on Radiation Patterns
The range of transmission phase that elements can achieve is a crucial factor that must be accounted for when selecting elements. It is often not possible to cover the entire 360° phase range using a conventional single patch of printed antenna elements, which usually cover around 240°. In such a case—when the transmission phase range is smaller than 360°—certain elements are likely to encounter unattainable phase shifts. Although the element selection process may minimize the impact of these errors by choosing the closest quantized values, these errors are fundamentally different from quantization errors and fall into the category of errors resulting from a limited phase range [20].
To better understand the influence of a limited phase range on the performance of a transmitarray antenna, the radiation patterns of the 13 × 13 aperture coupled transmitarray antenna based on Configurations A, B, and C considering different element phase ranges (240° and 360°) are presented in Fig. 9. It is well known that both the phase error and magnitude loss of transmitarray elements can increase SLL [20]. The results in Fig. 9 confirm that Configuration A, with a 240° phase shift, results in a higher SLL than Configurations B and C.
Simulated radiation pattern of the proposed transmitarray antenna for Configurations A, B, and C for 240° and 360° transmission phase ranges at a frequency of 10 GHz.
Furthermore, according to [23], lower spacing between unit cell elements can lead to lower SLL. The results in Fig. 9 also confirm this. Since Configuration C features lower spacing between patch elements of the array, it attained a lower SLL than Configuration B. Fig. 9 also confirms that Configuration C yields a lower SLL of −22 dB compared to the −19 dB and −15 dB observed for Configurations B and A, respectively.
2. Design of the MS-based Transmitarray Antenna
To expand the transmission phase range of the antenna, an MS sub-patch configuration (Configuration C) was employed. The previous section discussed various MS configurations in detail and determined the corresponding transmission phase ranges that could be achieved.
The transmitarray antenna, comprising 13 × 13 elements, was designed to operate at the center frequency of 10.5 GHz and featured an overall dimension of 192 mm × 192 mm (6.4λ0 × 6.4λ0, where λ0 is the free space wavelength at the center frequency). The transmitarray structure was fabricated on a Rogers Ro4003C substrate with a thickness of h = 0.813 mm. As mentioned earlier, to obtain the required phase shift for each unit cell element, the width of the MS, Wp, and length of the slot, Ls, were simultaneously varied according to Ls = Wp + 0.25 mm, thereby offering a very simple transmitarray design. The dimensions of each unit cell were carefully chosen to ensure that the desired distribution of phase shifts could be achieved across the entire array. The phase distribution is shown in Fig. 10. Photos of the corrugated conical horn antenna with a dielectric rod and the overall fabricated transmitarray antenna are presented in Fig. 11.
Compensated phase distribution.
3. Transmitarray Antenna Results
The feed horn and all the transmitarray elements were simulated using the Ansoft HFSS software package. The simulated normalized radiation patterns of the proposed antenna with configuration C at three different frequencies within the −1 dB gain limit are depicted in Fig. 12. Owing to the symmetry of the antenna structure, the radiation pattern exhibits very good symmetry. Furthermore, low cross-polarization is achieved due to the use of a rectangular slot in the middle plane.
Simulated E- and H-plane normalized radiation patterns of the proposed antenna with Configuration C MS unit cells at different frequencies: (a) 9.5 GHz, (b) 10.5 GHz, and (c) 11.5 GHz.
The radiation pattern of the fabricated transmitarray antenna with Configuration C MS unit cells at a center frequency of 10.5 GHz is illustrated in Fig. 13. The measured and simulated results show good agreement.
Measured (M) and simulated (S) normalized radiation patterns of the proposed antenna with Configuration C MS unit cell at a frequency of 10.5 GHz: (a) E-plane and (b) H-plane.
Fig. 14 presents the simulated and measured peak gain with regard to frequency. The results demonstrate that the transmitarray antenna achieved a −1 dB gain bandwidth of 2 GHz, which is approximately 19% of the central frequency at 10.5 GHz. Furthermore, the measured gain was approximately 1 dB lower than the simulated gain. This difference can be attributed to various factors related to the construction of the prototype, such as layer misalignment, the presence of an air gap between layers, misalignment of the phase center, tolerances in feed horn manufacturing, and manufacturing tolerances of the array. Fig. 14 also illustrates the variation in the aperture efficiency of the transmitarray antenna, indicating a peak value of almost 50%.
Measured and simulated peak gain and aperture efficiency of the proposed transmitarray antenna with Configuration C MS unit cells.
Table 3 presents a comparison of the proposed antenna with those previously reported in the literature [3, 13, 14, 24–32]. Compared to the complicated structures in [13] and [14] that employed aperture coupling mechanisms, the present work provides a similar performance while being much simpler in both design and fabrication, as well as lower in cost.
Comparison of features of the proposed transmitarray antenna with those of previously reported works
IV. Conclusion
In this paper, an MS patch is designed on a two-layer dielectric substrate with a slot on the ground plane between the layers. By implementing source-free characteristic mode analysis, it was found that MS patches can effectively operate in the wider mode. To verify the design strategy, the MS patches were combined with a coupling slot to realize a wideband unit cell. By changing the width of the MS patches and simultaneously tuning the slot length, a phase shift range of 0° to 360° was realized. The full-wave simulation and measured results demonstrate that the proposed transmitarray antenna is capable of realizing a −1 dB gain bandwidth of 19% and a maximum efficiency of 50%.
References
Biography
Masoumeh Darwish, https://orcid.org/0000-0001-9897-0205 received her B.Sc. and M.Sc. degrees in communication engineering from Shahed University, Tehran, Iran, in 2009 and 2011, respectively. She is currently working toward a Ph.D. degree in communication engineering at Shahed University, Tehran, Iran. Her main areas of interest are array antennas, wideband antennas, and microstrip antennas and components.
Hamid Reza Hassani, https://orcid.org/0000-0001-8272-6291 received his B.Sc. degree in communication engineering from Queen Mary College London, UK, in 1984, his M.Sc. degree in microwaves and modern optics from University College London, UK, in 1985, and his Ph.D. degree in microstrip antennas from the University of Essex, UK, in 1990. Currently, he is a professor in the Department of Electrical and Electronic Engineering, Shahed University, Tehran, Iran. His research interests include printed circuit antennas, phased array antennas, and numerical methods in electromagnetics.