Circularly polarized (CP) antennas are an attractive choice for wireless communication because of their reduced polarization loss factor and multi-path interference [1, 2]. Circular polarization is also a desirable feature in mobile devices for global positioning system applications [3, 4]. Moreover, we proposed a wideband CP monopole slot antenna using a square-shaped ground plane [5]. Electrically, small antennas of mobile devices suffer from several inherent problems, such as small bandwidth, high Q-factor, and low efficiency at smaller sizes [6, 7]. Therefore, small antennas are often employed to excite the large ground plane of a mobile device for effective radiation.

A loop-type ground radiation antenna (GradiAnt) showed good radiation performance in various studies [8–10]. The antenna’s performance is attributed to the antenna’s strong coupling with the ground plane. Enhancing the radiation efficiency of mobile antennas is a challenging task for antenna engineers. To enhance the radiation efficiency of the antennas of a mobile device that has an electrically small ground plane, ground mode tuning (GMT) is an effective technique [11, 12]. The GMT structures are installed at the edge of the ground plane to match the resonance frequency of the ground mode with the operating frequency of the antenna [13]. The theory of characteristic modes can be utilized in a systematic design of antennas. Application of the theory to the design of CP antennas has been proposed in [14]. The loop-type GradiAnts proposed in the literature are linearly polarized antennas. However, GradiAnts with circular polarization are not well documented in the literature.

In this paper, we propose a CP loop-type GradiAnt for Internet of Things (IoT) devices with compact-sized ground plane, operating at 2.45 GHz. To generate CP waves, two orthogonal current modes of the ground plane need to be excited by the antenna with a phase difference of 90° [15]. Implementing these conditions on a mobile device antenna while maintaining good radiation efficiency is a challenging task. In the antenna design process, we have employed the characteristic mode analysis of the proposed ground plane. Two orthogonally polarized modes with equal magnitude have been excited by designing the antenna at the corner of the ground plane. The GMT structure is utilized to tune one of the modes so that a phase difference of 90° is achieved between the two modes. The mode can be fine-tuned by using different values of inductor.

The loop-type GradiAnt acts as a magnetic coupler, and its coupling with the dominant ground mode is maximum when it is located at the maximum current location of the ground mode. Therefore, the optimum location of the antenna is at the middle of the edge of the ground plane. At this location, however, the antenna excites only one mode. To simultaneously excite two orthogonal modes on the ground plane with equal magnitude, the antenna should be located at the corner of the ground plane. At this location, the electric field of both modes is strongest; therefore, the electric coupling between the antenna and the ground modes should be enhanced. This can be expressed as [16]:

where Jⁱ is the impressed electric current density, E^g is the electric field of the ground plane, ω _o is the operating frequency of the antenna, and ω _g is the resonance frequency of the ground mode. The integration is carried over the volume of the antenna. Furthermore, the coupling between the GradiAnt and the ground mode can be enhanced by the optimum impedance level of the antenna’s resonance loop. The impedance level is expressed as (L _r /C _r )^1/2, where L _r is the inductance of the resonance loop and C _r is the resonance capacitor. The optimum impedance level depends on the antenna’s location on the ground plane and the size of the antenna’s resonance loop. If the antenna is located at the middle of the ground plane, the impedance level should be minimum, whereas if the antenna is located at the edge of the ground plane, the impedance level should be maximum [17]. The antenna’s impedance level can be enhanced by increasing the clearance area of the antenna and using a lower value of C _r . The impedance matching can be controlled by the feeding loop that contains C _f [18]. The impedance level of the feeding loop (L _f /C _f )^1/2 must be 50 Ω to match with the RF source. The operating frequency of the antenna is also determined by L _r and C _r . The geometry of the proposed antenna is shown in Fig. 1.

The antenna element is located at the left corner of the 45 mm × 45 mm ground plane by etching a square clearance of 7 mm × 7 mm in size. FR4 material (ɛ _r = 4.4, tanδ = 0.02) with 1 mm thickness is used as a substrate material. The resonance capacitor (C _r ) is located at the corner of the outer loop, called the resonance loop. The feeding loop is 4.5 mm × 4.5 mm in size and contains the feeding capacitor C _f . The GMT structure is placed at the bottom of the ground plane and consists of an inductor (L), a metallic strip 45 mm × 1 mm in size, and a shorting strip. The strip is oriented along the xz-plane, and the gap between the metallic strip and the ground plane is 2 mm. The strip is connected with the ground plane through the shorting strip and the inductor. The width of the shorting strip is 2 mm.

The resonance capacitor’s location on the resonance loop plays a critical role in the excitation of orthogonal modes on the ground [19]. When a low-valued C _r is located at P₁ on the resonance loop, high reactance of the capacitor can be modeled as an open circuit; thereby the GradiAnt acts as a monopole antenna attached to the location P₂. Therefore, according to Eq. (1), the antenna excites the current along the x-axis (mode 1) on the ground plane. Similarly, the antenna excites the current along the y-axis (mode 2) on the ground plane when C _r is located at P₂. Placing C _r at the corner of the resonance loop virtually divides the antenna into two monopoles attached at P₁ and P₂, respectively, resulting in the excitation of both modes with equal magnitude. Fig. 2 shows the simulated surface current density on the ground plane without the GMT structure. The simulated values of C _r and C _f were 0.15 pF and 0.12 pF, respectively, where the values have been optimized through full-wave simulations. It can be observed that the current is excited diagonally on the ground plane, which is the resultant direction of mode 1 and mode 2, demonstrating that both modes have been excited with equal magnitude.

To achieve circular polarization, the first two modes of the ground plane have been utilized, where ω _g of both modes is greater than ω ₀. The GMT structure is employed to decrease the resonance frequency of mode 2, so that a phase difference of 90° is achieved between modes 1 and 2 while maintaining equal magnitude in both modes. The resonance frequencies of the ground modes depend on the size of the ground plane; therefore, the antenna performance depends critically on the ground plane size. Simulations have been conducted to observe the effect of the size of the ground plane on antenna performance where the sizes of the antenna and the GMT strip have been unchanged. The observations are shown in Table 1. The data demonstrate that the increase in the ground plane size decreases the resonance frequency of both ground modes (f _g ), as well as the operating frequency of the antenna (f _o). The tabulated f _g is without GMT structure. The decrease in f _g is more significant as compared to f _o. The operating frequency can be tuned using C _r . Furthermore, it is observed that the antenna’s matching bandwidth (−6 dB ref.) increases with the increase in the ground size. According to Eq. (1), the coupling between the antenna and the ground mode increases if ω _g approaches ω _o . The higher coupling results in the increased matching bandwidth of the antenna. To decrease the resonance frequency of mode 2 of the ground plane, inductor (L) has been utilized, which appears in series with the ground mode. Therefore, increasing L decreases the resonance frequency of mode 2. Table 1 indicates that the lower values of L are used to achieve circular polarization with the increase in the ground size. Moreover, the axial ratio (AR) bandwidth increases with the increase in the ground size.

To validate the polarization of the antenna, the simulated surface current density of the proposed structure is presented in Fig. 3. Simulations show that the excited currents rotate in clockwise direction on the ground plane. Fig. 3(a) shows that the current density on the major portion of the ground plane is directed along the x-axis at a phase of 0°, i.e., mode 1 is dominant.

Similarly, Fig. 3(b) shows that, at the phase of 90°, the current is directed along the y-axis, i.e., mode 2 is dominant. It can be observed that the current density is stronger around the GradiAnt, showing that the antenna acts as an excitation element for the ground plane. Although the magnitude of current density around the antenna is strong, the current distribution on the ground plane produces effective radiation. Therefore, the time phase difference between both ground modes generates left-hand circularly polarized (LHCP) waves along the +z-axis and right-hand circularly polarized (RHCP) waves along the −z-axis.

The antenna was fabricated for experimental validation, as shown in Fig. 4. The reflection coefficient was measured using Agilent 8753ES vector network analyzer, and the radiation characteristics were measured using a 3D CTIA-OTA chamber. The measured and simulated reflection coefficients are presented in Fig. 5.

The simulated bandwidth of the antenna with reference to −6 dB was 100 MHz (2.4–2.5 GHz), whereas the measured bandwidth was 150 MHz (2.37–2.52 GHz), showing good agreement with the simulated result. The measured and simulated ARs in the direction of +z-axis are plotted in Fig. 6 along with measured total efficiency. The measured AR bandwidth with reference to 3 dB was 130 MHz (2.38–2.51 GHz) and the simulated AR was 90 MHz (2.41–2.5 GHz). The minimum value of measured AR was 1 dB at 2.44 GHz. The average efficiency of the antenna in the operating band is 65%, and the maximum efficiency of the antenna is 74% at 2.44 GHz. The efficiency of the antenna is suitable for wireless applications. Fig. 7 presents measured LHCP, RHCP, and peak gains as functions of frequency. In accordance with the measured efficiency, the maximum value of peak gain (1.66 dB) occurs at 2.45 GHz and decreases away from it. Measured LHCP and RHCP gains at the operating frequency are −0.36 dB and 0.64 dB, respectively. The normalized, simulated, and measured radiation patterns of the antenna at 2.45 GHz are displayed in Figs. 8 and 9, respectively. The RHCP and LHCP data in the xz- and the yz-planes are plotted in Fig. 8. The difference between simulated and measured results is mainly due to the feeding cable and the fabrication tolerance of the GMT structure. As shown, the higher values of LHCP and RHCP gain patterns occur in the upper and lower hemispheres, respectively, confirming that the antenna produces LHCP and RHCP waves along the +z- and −z-axes, respectively. In Fig. 8(a), it can be observed that the LHCP and RHCP patterns in the xz-plane are symmetric, whereas in the yz-plane the patterns are tilted towards −30° and −150°, respectively. The asymmetry is due to the asymmetric geometry of the proposed antenna where the GMT structure is installed in the yz-plane, whereas in the xz-plane there is no GMT structure. Moreover, as shown in Fig. 3, linear currents are excited in the GMT structure, causing the patterns to tilt in the yz-plane. The measured cross polarization levels of the antenna in +z and –z axes were below −25 dB. The simulated total gain radiation pattern is shown in Fig. 9(a). The pattern is isotropic in the xz-plane, whereas in the xy-plane, the minimum value of the pattern is −10.5 dBi, which occurs at 90°. Therefore, the antenna has a quasi-isotropic radiation pattern, suitable for wireless applications as compared to that of linearly polarized antennas having nulls in the radiation patterns [19]. A good agreement can be observed in the measured and simulated gain patterns in the xz- and yz-planes. The minimum value of the measured gain was −15.87 dBi, in the yz-plane. The agreement of the measured and simulated results verifies that the proposed antenna is suitable for IoT applications.

We proposed a CP loop-type ground radiation antenna using a ground mode tuning structure. The conditions of circular polarization were achieved successfully using the antenna design and ground mode tuning structure. The simulated and measured data showed good agreement. The antenna generated LHCP waves in the +z-axis and RHCP waves in the −z-axis with cross polarization levels less than −25 dB. The total gain radiation pattern of the antenna was quasi-isotropic, which is an attractive feature for internet of things applications. The matching bandwidth of the antenna was 150 MHz and the axial ratio bandwidth was 130 MHz, covering the complete 2.4–2.48 GHz band. The proposed technique is versatile and can be applied to other wireless applications as well.